Currently, there is a portable communication terminal (referred to as a mobile terminal in this specification hereafter) capable of corresponding to multiple wireless communication standards and multiple frequency bands. Corresponding to multiple standards is referred to as multimode correspondence, and corresponding to multiple frequency bands is referred to as multiband correspondence.
In recent years, a transmitter that also conducts frequency conversion to an RF transmission carrier frequency when converting a digital baseband signal to an analog signal (digital-to-analog conversion), and directly modulating a digital signal to an RF frequency is well known as a configuration of such a multimode/multiband corresponding terminal for transmission. Such a transmitter is disclosed in Patent Document 1, for example.
The invention disclosed in Patent Document 1 has an RF frequency converter that is similar to a Gilbert cell mixer incorporated to a part of series-connected transistors circuit for a well-known current control-type digital-to-analog converter. According to such a configuration, it is possible to have a digital-to-analog converter, an RF frequency converter or an RF modulator as independent circuits, and conduct digital-to-analog conversion and RF frequency modulation simultaneously and multi-functionally.
The transmitter disclosed in Patent Document 1 can be called a digital-to-RF converter, a direct RF converter, or a direct RF modulation transmitter constituted thereby. It has several merits such as being able to omit an analog baseband filter circuit between the digital-to-analog converter and the RF frequency converter, which is normally required for the conventional transmitter that carries out separate operations.
FIG. 6 illustrates the configuration of the direct RF modulation transmitter described above. The direct RF modulation transmitter illustrated in FIG. 6 is constituted by two digital-to-RF converters (DRC) 1 and 2, a Divide-by-2 divider 3, and an output matching circuit 4.
An RF signal Loin+ for multiplying frequencies (referred to as transmission local RF signal hereafter) and a transmission local RF signal Loin−, which results from inverting the phase of the RF signal Loin−, are externally supplied to the Divide-by-2 divider 3. The Divide-by-2 divider 3 receives the transmission local RF signals Loin+ and Loin−, generates two pairs of differential local signals TxLoI+ and TxLoI−, and TxLoQ− and TxLoQ− each differing in phase by 90 degrees, and outputs them to the DRCs 1 and 2. In this example, since differential local signals of 0 degrees and 90 degrees are generated by the Divide-by-2 divider 3, frequency of the transmission local RF signals Loin+ and Loin− is double the target frequency of a transmission carrier wave. The frequency of the differential local signals TxLoI+, TxLoI−, TxLoQ+, and TxLoQ− is that of the transmission carrier wave. There is a 90 degree phase difference between the differential local signals TxLoI+ and TxLoI−, and TxLoQ+ and TxLoQ−.
The DRC 1 and the DRC 2 have the same configuration. The differential local signals TxLoI+ and TxLoI−, and TxLoQ+ and TxLoQ− are supplied to the DRC 1 and the DRC 2 with the same type of phase difference as that of a so-called IQ direct modulator, thereby constituting the direct RF modulation transmitter. That is, an I (In-phase) digital baseband signal (referred to as ‘IBB Data’ in the drawing) is input to the DRC 1. Moreover, a Q (Quadrature) digital baseband signal (referred to as ‘QBB Data’ in the drawing) is input to the DRC 2.
Furthermore, a sampling clock signal CLKBB is input to the DRCs 1 and 2. The DRCs 1 and 2 are signal converters, each having an integrated function of a digital-to-analog converting function and a frequency multiplying function for converting the frequency of a baseband signal to the RF signal. Through such functions, the DRC 1 outputs an output differential signal based on the clock signal CLKBB, the I digital baseband signal, and the differential local signal. Moreover, the DRC 2 outputs an output differential signal based on the clock signal CLKBB, the Q digital baseband signal, and the differential local signal. The output differential signals from the DRCs 1 and 2 are added together, and the resulting signal is output as a carrier wave via the output matching circuit 4 and a power amplifier (referred to as PA in the drawing) in the subsequent stage.
The output matching circuit 4 is a circuit that is constituted by passive elements, such as a capacity and/or an inductor element, and has a bandpass-type gain characteristic, which has the frequency of the transmission carrier wave as a central frequency. Note that the direct RF modulation transmitter illustrated in FIG. 6 carries out addition of the output differential signal output from the DRC 1 and the output differential signal output from the DRC 2 through direct connection of signal paths on the premise that the DRCs 1 and 2 output currents.
FIG. 7 illustrates a circuit having the configuration of each of the DRC 1 and the DRC 2 disclosed in the aforementioned Patent Document 1. Each of the DRC 1 and the DRC 2 includes a block for processing signals on the least significant bit (LSB) side, and a block for processing signals on the most significant bit (MSB) side. The block on the LSB side is constituted by current sources 200, 201, . . . 20k, in each of which unit cells are weighted in a binary manner, local signal switches 220, 221, . . . 22k arranged in a Gilbert cell form, and data signal switches 240, 241, . . . 24k. 
The block on the MSB side has a structure in which current sources 210, each weighted by the same value, and local signal switches 230 and data signal switches 250 arranged in a Gilbert cell form are respectively connected in parallel where the number of respective components is equal to the number of required bits. According to such a structure, the direct RF modulation transmitter disclosed in Patent Document 1 can conduct digital-to-analog conversion and frequency multiplication simultaneously. Note that in the example illustrated in FIG. 7, current outputs from all cells are converted to respective voltages by external loads deployed outside of the DRCs.
FIG. 8 is a diagram for describing a typical operation of a circuit called digital-to-RF converter or direct RF converter. Such a circuit receives an RF signal and a digital baseband signal and modulates the RF signal in accordance to the digital baseband signal and then outputs the resulting modulated signal. Regarding the modulated signal, a phase inverted signal of the transmission carrier wave is output at the time when the digital baseband signal is changed over.
Noise of an output signal from the direct RF modulation transmitter will now be described. With the direct RF modulation transmitter, the main factors for determining a noise floor near the carrier wave of the output signal are Thermal noise and Flicker noise, which generates from an inner element, and quantization noise, which generates in the digital-to-analog conversion process. A transmitter for conducting digital-to-analog conversion and frequency multiplication with separate circuit blocks allows installation of an analog filter immediately after digital-to-analog conversion. Therefore, hardly any quantization noise is included in the signal after frequency multiplication.
Whereas the conventional DRC illustrated in FIG. 7 has an integrated function of a digital-to-analog converting function and a frequency multiplying function, as mentioned above. Therefore, the quantization noise generated during digital-to-analog conversion is output as noise near the carrier wave. As a result, with the conventional DRC illustrated in FIG. 7, generation of quantization noise during digital-to-analog conversion needs to be controlled to a minimum.
The following Equation 1 represents quantization noise amount generating during digital-to-analog conversion when a typical digital-to-analog converter has output a desired full scale wave signal. Equation 1 represents noise amount when a desired wave signal level is made as a reference, wherein B denotes bit number, and fs denotes sampling frequency.NoiseFloor(dBc/Hz)=−{6/02·B+1.76+10·log(fs/2)}  Equation 1
Equation 2 represents quantization noise amount in the case where a digital-to-analog converted signal undergoes frequency multiplication so as to be converted to a higher-frequency wave when the DRC illustrated in FIG. 7 has output a desired full scale wave signal. According to Equation 1 and Equation 2, it is understood that increasing either the bit number B or the sampling frequency fs is necessary for reducing the noise. When considering implementation of low quantization noise in a complementary metal oxide semiconductor (CMOS) circuit, the maximum implementable frequency should be set as the sampling frequency, and insufficient reduction of noise should be compensated by increase in bit number.NoiseFloor(dBc/Hz)=−{6.02·B+1.76+10·log(fs/2)+10·log(2)}  Equation 2
If the DRCs 1 and 2 illustrated in FIG. 7 are made up of MOS transistors, current sources 200 to 20k and 210 occupy most of the area of the DRCs 1 and 2. The area of the current sources 200 to 20k and 210 is determined with accuracy of fluctuation in current, which is calculated from bit number of an input digital signal and required linearity (distortion characteristic). The bit number of the input digital signal and the required linearity depend on the target quantization noise level of the direct RF modulation transmitter.
Relative fluctuation of the current output from the MOS transistors is given in Equation 3. σI/I in Equation 3 denotes standard deviation of the relative fluctuation of current. Aβ and AVT denote parameters for fluctuation dependant on the semiconductor process, VGS denotes a voltage between a gate and a source of the MOS transistor, Vt denotes a threshold voltage of the MOS transistor, W denotes channel width of the MOS transistor, and L denotes channel length of the MOS transistor.
                              (                                    σ              I                        I                    )                =                                                            1                2                            ⁡                              [                                                      A                    β                    2                                    +                                                            4                      ⁢                                              A                        VT                        2                                                                                                            (                                                                              V                            GS                                                    -                                                      V                            T                                                                          )                                            2                                                                      ]                                                                    (                WL                )                            min                                                          Equation        ⁢                                  ⁢        3            
In the case where the bit number of the digital signal to be converted has increased, maintaining quality of the linearity equal before and after increase is considered. If making a required value for relative fluctuation of current be ½1/2 is taken into consideration, according to the aforementioned Equation 3, it is necessary to double the area occupied by the current sources by increasing one bit of the digital signal. Moreover, as the required element number for the configuration of the DRCs 1 and 2 is doubled by increasing one bit, the current source area quadruples overall. As a result, the method for increasing bit number to reduce the quantization noise has a drawback of an increase in area of the DRCs 1 and 2.
Furthermore, the RF transmitter for a wireless communication device is not required to output a uniform amount of noise included in an output RF signal across the entire frequency band, but frequency bands with strict noise requirements and frequency bands without strict requirements are mixed therein. For example, an RF transmitter based on W-CDMA, which is a cellular phone standard, is applied to a frequency division duplex (FDD) system in which reception and transmission are conducted simultaneously, wherein noise requirement near a reception frequency is most strict.